AN SDR-BASED FRS TRANSCEIVER
PROTOTYPING A MULTI-CHANNEL FRS TRANSCEIVER WITH CTCSS
SUPPORT VIA GNU RADIO AND THE NU AND BLADERF
JUNE 21, 2016
An SDR-based FRS Transceiver Nuand, LLC
License
This work by Nuand, LLC is licensed under:
Creative Commons Attribution 4.0 International License
Authors
Robert Ghilduta Brian Padalino
<robert.ghilduta@nuand.com> <brian.padalino@nuand.com>
Nuand, LLC Nuand, LLC
Jon Szymaniak
<jon.szymaniak@nuand.com>
Nuand, LLC
Contributing Authors
Dr. Juan R. Pimentel
<jpimen[email protected]>
Professor of Computer Engineering
Kettering University
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An SDR-based FRS Transceiver Nuand, LLC
Revisions
Comments, feedback, improvements, and fixes may be sent to <bladeRF@nuand.com>.
Revision Date Summary
1 2015-06-12 Initial public draft
2 2015-10-25 Incorporated feedback and revisions from Dr. Juan R. Pimentel
3 2016-01-07 Fixed incorrect frequency values in Figure 1
4 2016-06-21 Typo fix in Section 5.4
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An SDR-based FRS Transceiver Nuand, LLC
Contents
1 Intended Audience 1
2 Abstract 1
3 Overview of the Family Radio Service 2
4 Transceiver Design 3
4.1 Transmit Path . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
4.1.1 Audio Input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
4.1.2 Optional CTCSS Tone . . . . . . . . . . . . . . . . . . . . . . . . . . 6
4.1.3 FM Modulation and Mute Blocks . . . . . . . . . . . . . . . . . . . . 7
4.1.4 Polyphase Synthesizer . . . . . . . . . . . . . . . . . . . . . . . . . . 8
4.1.5 Post-PS gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
4.1.6 Interpolating Filters . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
4.1.7 Mixing and Hardware Sink . . . . . . . . . . . . . . . . . . . . . . . . 12
4.2 Receive Path . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
4.2.1 Hardware Source and Valve . . . . . . . . . . . . . . . . . . . . . . . 14
4.2.2 Power Meter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
4.2.3 Frequency Translation and Decimating FIR Filters . . . . . . . . . . 15
4.2.4 Polyphase Channelizer . . . . . . . . . . . . . . . . . . . . . . . . . . 16
4.2.5 Squelch and FM Demodulation . . . . . . . . . . . . . . . . . . . . . 17
4.2.6 CTCSS Tone IIR Filter . . . . . . . . . . . . . . . . . . . . . . . . . . 19
4.2.7 Audio Sinks . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
5 Testing and Evaluation 21
5.1 Flowgraph GUI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
5.2 Nuand bladeRF SDR . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
5.3 Reception . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
5.4 Transmission . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
6 Software and System Configuration 29
7 Conclusions 29
8 Flowgraph Diagrams 30
References 33
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An SDR-based FRS Transceiver Nuand, LLC
List of Tables
1 FRS frequency allocations . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
2 Common CTCSS tone frequencies, in Hz . . . . . . . . . . . . . . . . . . . . 2
3 Filter parameters of TX audio LPF . . . . . . . . . . . . . . . . . . . . . . . 7
4 Filter parameters for polyphase synthesizer LPF . . . . . . . . . . . . . . . . 9
5 Filter parameters for first TX interpolating (5x) FIR filter . . . . . . . . . . 11
6 Filter parameters for second TX interpolating (4x) FIR filter . . . . . . . . . 12
7 Filter parameters for RX frequency translating and decimating FIR filter . . 15
8 Filter parameters for RX decimating FIR filter . . . . . . . . . . . . . . . . . 16
9 Filter parameters of RX CTCSS-blocking high pass filter . . . . . . . . . . . 19
10 Minimum required component versions . . . . . . . . . . . . . . . . . . . . . 29
List of Figures
1 Transceiver tuning with respect to FRS channel allocations . . . . . . . . . . 4
2 Avoiding effect of IQ imbalance via offset tuning . . . . . . . . . . . . . . . . 4
3 MATLAB Filter Design and Analysis Tool . . . . . . . . . . . . . . . . . . . 5
4 FRS TX block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
5 Magnitude response of TX audio LPF . . . . . . . . . . . . . . . . . . . . . . 7
6 Polyphase Synthesizer configuration for FRS channels 1-7 . . . . . . . . . . . 8
7 Magnitude response of polyphase synthesizer FIR filter . . . . . . . . . . . . 10
8 Magnitude response TX interpolating FIR filters . . . . . . . . . . . . . . . . 12
9 FRS RX block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
10 Magnitude response of RX decimating FIR filters . . . . . . . . . . . . . . . 16
11 Magnitude response of NBFM de-emphasis filter . . . . . . . . . . . . . . . . 18
12 Response of RX CTCSS-blocking high pass filter . . . . . . . . . . . . . . . . 20
13 RX Section of the flowgraph GUI . . . . . . . . . . . . . . . . . . . . . . . . 22
14 TX Section of the flowgraph GUI . . . . . . . . . . . . . . . . . . . . . . . . 22
15 Nuand bladeRF architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
16 Lime Microsystems LMS6002D architecture . . . . . . . . . . . . . . . . . . 24
17 Cobra CXT545 FRS handheld transceiver used to test flowgraph . . . . . . . 25
18 FFT plot of digital signal containing FRS channels 8-14 . . . . . . . . . . . . 26
19 FFT plot of digital signal containing FRS channel 8 . . . . . . . . . . . . . . 27
20 FFT plot of digital signal containing all FRS channels . . . . . . . . . . . . . 27
21 FFT plot of analog signal containing FRS channels 1-7 . . . . . . . . . . . . 28
22 FFT plot of analog signal containing all 14 FRS channels . . . . . . . . . . . 28
23 Full FRS transceiver flowgraph . . . . . . . . . . . . . . . . . . . . . . . . . 30
24 RX path of FRS transceiver . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
25 TX path of FRS transceiver, slightly re-arranged to fit page . . . . . . . . . 32
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List of Acronyms and Abbreviations
CTCSS Continuous Tone-Coded Squelch System
DAC Digital to Analog Converter
FIR Finite Impulse Response
FPGA Field Programmable Gate Array
FRS Family Radio Service
IIR Infinite Impulse Response
GRC GNU Radio Companion
GUI Graphical User Interface
LNA Low Noise Amplifier
LPF Low Pass Filter
RX Receive
SINAD Signal-to-Noise and Distoration ratio
SDR Software Defined Radio
TX Transmit
VGA Variable Gain Amplifier
VSA Vector Signal Analyzer
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An SDR-based FRS Transceiver Nuand, LLC
Intended Audience
The intended audience for this document is someone with a basic understanding of fre-
quency domain analysis, sampling theory, digital filtering, and analog modulation techniques.
Prior experience with GNU Radio, GNU Radio Companion (GRC), Software Defined Ra-
dio (SDR), and MATLAB is also beneficial. This paper is likely to be useful for SDR users
looking for examples of:
Narrow-band FM transmitters and receivers
Filter design using software tools
Usage of GNU Radio’s polyphase channelizer and synthesizer blocks
Full-duplex operation of the bladeRF
Abstract
The Nuand bladeRF is a USB 3.0 SDR featuring full-duplex operation and configurable
filter bandwidths that can cover up to 28 MHz. These properties are leveraged to design and
implement a real-time Family Radio Service (FRS) transceiver capable of simultaneously
transmitting and/or receiving on any subset of its 14 channels, with optional Continuous
Tone-Coded Squelch System (CTCSS) support. The open source GNU Radio framework
and the GRC tool are used to realize the transceiver design as a graphical flowgraph and
to create an interactive GUI interface for the transceiver. This paper presents both a high-
level description of the design, as well as a detailed discussion of the GNU Radio blocks
utilized and the rationale for their associated parameter values. In doing so, considerations
for targeting the bladeRF and accounting for real-world nonidealties are included. Lastly,
the approaches for testing and evaluation the design using RF test equipment and FRS
handhelds is presented, along with results.
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An SDR-based FRS Transceiver Nuand, LLC
Overview of the Family Radio Service
FRS is a private, 14-channel, two-way data communications service intended for short-
distance voice and data communications [1]. Rules governing this service are defined in
Title 47, Part 95B [2].
The frequency allocations for each of the 14 channels are listed below. Each channel is
allocated 12.5 kHz of bandwidth, and the max deviation of the narrow band FM signals
used on these channels is 2.5kHz [2].
Table 1: FRS frequency allocations
Channel Frequency (MHz) Channel Frequency (MHz)
1 462.5625 8 467.5625
2 462.5875 9 467.5875
3 462.6125 10 467.6125
4 462.6375 11 467.6375
5 462.6625 12 467.6624
6 462.6875 13 467.6875
7 462.7125 14 467.7125
Many handheld FRS radios include CTCSS support
1
, in which receivers require the presence
of a sub-audible tone in order to break squelch. Table 2 presents commonly used frequencies
for these sub-audible tones. Note that these CTCSS tones are not part of the FRS standard,
but an industry-created and adopted method for co-channel use between multiple parties.
Table 2: Common CTCSS tone frequencies, in Hz
67.0 71.9 74.4 77.0 79.7 82.5 85.4 88.5
91.5 94.8 97.4 100.0 103.5 107.2 110.9 114.8
118.8 123.0 127.3 131.8 136.5 141.3 146.2 151.4
156.7 162.2 167.9 173.8 179.9 186.2 192.8 203.5
210.7 218.1 225.7 233.6 241.8 250.3
1
CTCSS support is often marketed as “sub-codes” or “privacy codes.”
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An SDR-based FRS Transceiver Nuand, LLC
Transceiver Design
The transceiver presented in this document is designed to simultaneously receive and trans-
mit on any subset of the 14 FRS channels using GNU Radio [3] and the Nuand bladeRF SDR
[4]. This section first presents some of the driving factors of the design, followed by a detailed
description of both the Transmit (TX) and Receive (RX) portions of the flowgraph. It is
recommend that one have the flowgraph (or the screenshots in Section 8) readily accessible
while reading this section.
Although the individual FRS channels have a very small bandwidth, there is a 4.85 MHz
separation between channels 7 and 8. Instead of re-tuning the transceiver to the desired
frequency, the bladeRF is tuned to an intermediate frequency and its bandwidth and sample
rate are configured to cover the entire FRS spectrum. As the bladeRF is capable of operating
on up to 28 MHz of bandwidth, this does not pose a significant challenge.
This approach has the following advantages:
No time is spent re-tuning or sweeping across channels.
Calibration only needs to be performed for the intermediate frequencies used.
The tuning of the RX and TX channels in the context of the FRS spectrum is presented in
Figure 1. Note that both the TX and RX center frequencies are not in the “center” of the
FRS spectrum, as also shown in Figure 2(a). If the transceiver were tuned directly to the
“center” of the FRS spectrum, any IQ imbalance that remains after calibration could cause
mirror images of channels 1-7 to interfere with channels 8-14, and vice versa. This situation
is depicted in Figure 2(b). To avoid this, the RX and TX channels are tuned at 200 kHz
offsets from the “center” of the FRS spectrum, which is illustrated in Figure 2(c).
It is important to be aware of some trade-offs with this design, which must be be kept in
mind during the implementation and evaluation of the transceiver:
1. Using over 5 MHz of bandwidth concurrently on TX and RX will exceed USB 2.0
capabilities. A USB 3.0 connection is required.
2. By not tuning hardware to one of the FRS channels, any LO leakage from the trans-
mitter may interfere with other signals.
3. The offset tuning approach places any IQ images out-of-band, which may interfere with
neighboring signals.
We argue that #1 is reasonable, given the ever increasing availability of USB 3.x controllers
on modern systems.
Trade-offs #2 and #3 include interference and out-of-band emissions and are therefore serious
concerns. Fortunately, both DC offset and IQ imbalance can be compensated for using
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1 2 3 4 5 6 7
462.5625 MHz
462.7125 MHz
8 9 10 11 12 13 14
467.5625 MHz
467.7125 MHz
465.1375 MHz
TX
464.9375 MHz
RX
465.3375 MHz
2.3 MHz
2.7 MHz 2.3 MHz
2.7 MHz
Figure 1: Transceiver tuning with respect to FRS channel allocations
Ch 1-7
Ch 8-15
(a)
Ch 1-7
Ch 8-15
(b)
Ch 8-15
Ch 1-7
Ch 1-7
Ch 8-15
(c)
Ch 8-15
Ch 1-7
(a) Nominal case: Tuning to “center” and no IQ imbalance.
(b) Realistic case: IQ imbalance yields mirror images interfering with signals.
(c) Offset tuning to ensure IQ imbalance images do not overlap with signals.
Figure 2: Avoiding effect of IQ imbalance via offset tuning
hardware features of the bladeRF. In evaluating this design, these signals will be important
to measure and take note of.
In designing with this approach, we must first determine how the transmitter and receiver
will utilize the spectrum. Given the 25 kHz separation between channels and each channel’s
12.5 kHz of bandwidth, this implies that the bladeRF needs to be configured for a minimum
bandwidth defined by the following:
BW
TX, min
= 2 × (Max (|F
T X
− F
CH 1
|, |F
T X
− F
CH 14
|) + 6.25 kHZ)
= 2 × (Max (|464.9375 − 462.5625|, |464.9375 − 467.7125|) MHz + 6.25 kHZ)
= 5.5625 MHz
The equation for BW
RX, min
is similar, but instead uses an (F
RX
= 465.1175 MHz) term.
The Lime Microsystems LMS6002D transceiver [5] on the bladeRF has discrete TX Low Pass
Filter (LPF) bandwidth settings, with the next largest value being 6.0 MHz in quadrature,
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which passes [−3 MHz, 3 MHz] at baseband. The rejection of the LMS6002D 3 MHz filter
exceeds 50 dB at approximately ± 4 MHz [6]. On RX, this filter is used as an anti-aliasing
filter to ensure incoming signals outside of the sampling bandwidth are sufficiently attenuated
before being converted by the ADC. On TX, this filter is used to remove the images created
due to the zero-order hold nature of the DAC. Therefore, to avoid these artifacts, an 8 MHz
sample rate is used by the TX and RX modules.
These sample rate and bandwidth calculations provide constraints which guide the design of
the TX and RX paths detailed in the following subsections.
MATLAB’s fdatool, shown in Figure 3 is used to design the digital filters presented in
this document. However, GNU Radio’s gr filter design program and/or Octave could
also be used.
Figure 3: MATLAB Filter Design and Analysis Tool
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Transmit Path
Audio Input
At the beginning of the TX path, audio enters through an Audio Source block. Through
the use of the “pulse” argument provided to this block, the audio from the host machine’s
PulseAudio [7] subsystem is used. A 25 kHz sample rate is selected since it shares all common
factors with the final DAC output sample rate of 8 MHz. This alleviates the need for digital
resampling in the flowgraph prior to entering a downstream Polyphase Synthesizer
block.
bladeRF Sink
8 Msps
Interpolating FIR
↑4
Interpolating FIR
↑5
Audio Source
25kHz
Audio LPF
CTCSS Tone
Source
+
NBFM Mod
Polyphase
Synthesizer
↑16
X
Upper LO Offset
Source
Interpolating FIR
↑4
Interpolating FIR
↑5
Polyphase
Synthesizer
↑16
X
Lower LO Offset
Source
+
Figure 4: FRS TX block diagram
A Valve block is used to start and stop the flow of samples to the downstream portions of the
flowgraph. This value is “closed” when the user presses the “Push to Talk” Graphical User
Interface (GUI) button or checks the “Continuous TX” checkbox. When the valve is opened,
downstream blocks are not processing samples which saves CPU cycles when the transmit
functionality is not enabled. When no channels are selected to transmit, the bladeRF’s TX
front end is turned off. This helps guarantee no residual carrier is being transmitted when
there is nothing to transmit.
A Finite Impulse Response (FIR) Filter block is used to realize an LPF to limit
the audio frequency response to 3.125 kHz [2]. Using fdatool and the parameters shown in
Table 3, a filter whose magnitude response is shown in Figure 5 is created. From this filter’s
magnitude response, is is clear that frequencies above the 3.125 kHz limit are attenuated by
60 dB, while the vast majority of the human vocal range is passed.
Optional CTCSS Tone
Next, a Selector block is used to pass either the audio, or the audio added with a sub-
audible CTCSS tone supplied by a Signal Source block, to the next stage. The frequency
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Table 3: Filter parameters of TX audio LPF
Design Method FIR: Equiripple
Density Factor 20
Order 74
Sampling Frequency (kHz) 25
Passband Frequency (kHz) 2
Stopband Frequency (kHz) 3.125
Passband Weight 1
Stopband Weight 1
Frequency (kHz)
0 2 4 6 8 10 12
Magnitude (dB)
-70
-60
-50
-40
-30
-20
-10
0
Figure 5: Magnitude response of TX audio LPF
of the CTCSS tone is controlled by a drop-down widget in the GUI. When “None” is se-
lected, the selector passes the audio from the Audio Source block. Otherwise, the Signal
Source block is configured for the desired CTCSS frequency, and the Selector passes
the combination of the audio and this tone.
FM Modulation and Mute Blocks
The output of the Selector is fed into the NBFM Transmit block. There is no need to
interpolate here, so the Quadrature Rate is left the same as the sample rate of the incoming
audio.
No pre-emphasis is desired. As of GNU Radio commit ee369b92 (2015-03-04), the NBFM
Transmitter block does not currently implement pre-emphasis
2
, so the tau parameter is
2
The IIR filter is constructed with hard-coded feed-forward and feed-back taps set to 1.0.
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unused. However, this tau value is used simply to match the value used on the RX side of
the flowgraph, should this change in the future. The rationale for this value is discussed in
Section 4.2.5.
The FM modulated signal is then fed into a series of Mute blocks. Each of these blocks is
associated with one of the FRS channels, and is controlled by a GUI checkbox. When the
channel is not muted, the FM modulated signal is passed to the next stage. When muted,
values of (0 + 0j) are supplied to the next stage. Note the important difference here between
the Mute and Valve blocks – the former zeroes samples and keeps the stream running,
whereas the latter stops the flow of samples entirely. Using a Valve here would cause the
following block to stop running if one or more channels were muted, which is undesirable.
Polyphase Synthesizer
Because the FRS channels within their groupings are evenly spaced, GNU Radio’s Polyphase
Synthesizer [8] provides a very simple solution for combining multiple channels into a
single interpolated signal.
Two Polyphase Synthesizer blocks are used in conjunction with each other to con-
struct two 400 kHz wide signals – one containing FRS channels 1-7, and a second 400 kHz
wide signal containing FRS channels 8-14. Channels 4 and 11 are located at the center fre-
quencies of their respective signals. Each channel in the output is separated by the desired
25 kHz, as this is the input sample rate.
As noted in the previous section, input channels contain either the FM modulated audio
signal, or (0 + 0j), depending on the state of the associated mute block.
(8, N/A)
Legend: (Polyphase Synthesizer Channel, FRS Channel Number)
(0, 4)
(1, 5)
(2, 6)
(3, 7)
(13, 1)
(14, 2)
(15, 3)
0 kHz-200 kHz +200 kHz
(4, N/A)
(5, N/A)
(6, N/A)
(7, N/A)
(8, N/A)
(9, N/A)
(10, N/A)
(11, N/A)
(12, N/A)
Figure 6: Polyphase Synthesizer configuration for FRS channels 1-7
Although only 7 of these 25 kHz channels are required, these blocks are configured for a
total of 16 channels, with 9 of the channels left unused. This is done to alleviate the need
for more interpolation in later blocks; a value of “16” is selected to avoid necessitating an
additional factor of 2 in later interpolating blocks. The extra unused channels allow for a
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An SDR-based FRS Transceiver Nuand, LLC
wider transition band to be used in the interpolating FIR filter that follows this stage, which
reduces the required number of filter taps
3
.
The Channel Map property is set to [13, 14, 15, 0, 1, 2, 3], in order to position
FRS channels 1-7 (on input indices 0 through 6) as shown in Figure 6. With respect to
channel mapping, the same approach is taken for FRS channels 8 through 14 in the second
Polyphase Synthesizer block.
To minimize the impact of aliases, the Polyphase Synthesizer applies a per-channel
filter at the output sample rate of 400 kHz. The filter used in this block passes signals within
the 12.5 kHz FRS bandwidth limit. Therefore, the pass band is located at
12.5
2
kHz. To reject
the image occurring at the next channel, the stop band is place before the “edge” of the
image, expected to occur at 18.75 kHz = 25 kHz −
12.5
2
kHz.
The parameters used with fdatool to generate these filter taps are shown in Table 4. As
shown by the magnitude response in Figure 7, spectral content outside of the desired region
should be significantly rejected. However, one must take note of this undesired spectral
content when evaluating the design and measuring out-of-band emissions.
Table 4: Filter parameters for polyphase synthesizer LPF
Design Method FIR: Equiripple
Density Factor 20
Order 191
Sampling Frequency (MHz) 400
Passband Frequency (kHz) 6.25
Stopband Frequency (kHz) 18.75
Passband Weight 1
Stopband Weight 1
Post-PS gain
A Multiply Const block follows each Polyphase Synthesizer. These apply gain
to compensate for the reduction of average power resulting from zero-stuffing during inter-
polation. However, the gain applied here is not a constant value. Instead, it is a function
of the number of enabled channels that are being supplied to the input of the Polyphase
Synthesizer. The goal here is to apply gain to the signal such that the output IQ values
remain within [-1.0, 1.0].
With only one input enabled, a gain that is just slightly less than the interpolation factor is
used to avoid clipping. Thus, a value of 16×0.95 is used. When a second input is enabled, the
system is combining two signal sources whose amplitudes may be within [0.0, 1.0], yielding
a signal with an amplitude within [0.0, 2.0]. Therefore, to keep our signal amplitude within
3
This reduces the number of convolution operations required.
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Figure 7: Magnitude response of polyphase synthesizer FIR filter
the desired range, a division by two is applied, yielding a gain factor of
16×0.95
2
. By carrying
this logic through for our total of 7 channels, one arrives at the following gain expression
used for channels 8-15, with a similar expression being used for channels 1-7.
1 if tx num enabled ch8 14 == 0 else 16
*
0.95 / tx num enabled ch8 14
tx num enabled ch8 14 is a variable that represents how many channels are currently
enabled, based upon the state of GUI checkboxes. If no channels are enabled, a value of one
is used as a default. Note that this expression is written such that the check for this zero
case is first. This is done to ensure the logic short circuit does not cause a division by zero
to be evaluated.
Technically, the gain could be applied by scaling the tx pfs taps filter taps by this desired
gain value. This would have the advantage of removing extraneous multiplications from the
system. However, since these blocks are operating at a lower sample rate (400 kHz), they
have been left here to serve as an explicit reminder of this scaling operation.
The outputs of these blocks are connected to Qt GUI Waterfall Sink blocks to provide
feedback to the user that the channels selected via checkboxes are indeed being transmitted
on, as well as to provide visibility of the signal power and bandwidth.
Interpolating Filters
Ultimately, the two groups of FRS signals must be recombined into a single signal that
has been interpolated by a factor of 20 to match the hardware’s 8 Msps sample rate. This
interpolation is achieved utilizing two Interpolating FIR Filter blocks. As indicated
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by the block name, these first interpolate by zero-stuffing samples and then filter to reject
image artifacts that result from zero-stuffing.
The first FIR filter interpolates by a factor of 5, yielding an output sample rate of 2 MHz.
The second stage interpolates by a factor of 4, yielding the final output sample rate of 8
MHz.
The parameters for these filters are shown in Tables 5 and 6.
To account for energy lost due to the interpolation, the filter taps obtained via fdatool are
scaled by their associated interpolation factor. Figure 8 presents the magnitude responses
of the individual stages, as well the combined response of both stages.
The passband frequencies are selected to ensure all 7 of the 25 kHz channels are passed
without attenuation. Of the 400 kHz of bandwidth provided to the input of the first filter,
175 kHz
4
may contain information. Therefore, we set our passband to 85.7 kHZ =
175
2
kHz.
The stop band for the first filter is placed at the location where the first image artifact caused
by interpolation is expected to be. This is calculated as follows:
F
stop
=
F
s
2
+
F
s
2
− BW
= 200 kHz + (200 kHz − 87.5 kHz)
= 312.5 kHz
The position of the second filter’s stop band is selected such that the first null of this filter
aligns with null occurring before the image of the previous filter’s response. This can be
quickly achieved through an iterative process of measuring the location of the first null,
changing the F
stop
value in fdatool, and repeating until the nulls align sufficiently.
Table 5: Filter parameters for first TX interpolating (5x) FIR filter
Design Method FIR: Equiripple
Density Factor 20
Order 59
Sampling Frequency (MHz) 2
Passband Frequency (kHz) 87.5
Stopband Frequency (kHz) 312.5
Passband Weight 1
Stopband Weight 1
The motivation behind using two stages rather than one is to minimize the number of
convolution operations performed by the filter per unit of time. If a single interpolation by
4
Recall that seven 25 kHz inputs are provided to the Polyphase Synthesizer.
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Table 6: Filter parameters for second TX interpolating (4x) FIR filter
Design Method FIR: Equiripple
Density Factor 20
Order 27
Sampling Frequency (MHz) 8
Passband Frequency (kHz) 87.5
Stopband Frequency (MHz) 1.67
Passband Weight 1
Stopband Weight 1
Frequency (MHz)
0 0.5 1 1.5 2 2.5 3 3.5
Magnitude (dB)
-200
-150
-100
-50
0
Combined Response
Second (4x) Stage
First (5x) Stage
Figure 8: Magnitude response TX interpolating FIR filters
20 were performed, all of the filtering would be done at the final 8 MHz samplerate. Because
the FRS spectral content is in a relatively small percentage of the interpolated signal, this
requires a large number of taps to achieve the desired transition band and rejection. This
combination of higher sample rate and more taps leads to higher computational load. In
general, by performing the interpolation and filtering in stages, it is possible to achieve fewer
operations per stage, with each stage running at a fraction of the final sample rate.
Mixing and Hardware Sink
After the previous stages of interpolation, there is ample bandwidth to combine the two
groups of 7 FRS channels. Each group is mixed from baseband to a target frequency offset,
per Figure 1, and then added to form the final output signal. This mixing is performed
though multiplication with the output of a Signal Source block.
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An SDR-based FRS Transceiver Nuand, LLC
The Signal source is configured to provide a cosine at the target offset frequency (-2.3
MHz for Ch 1-7 and 2.7 MHz for Ch 8-14). The amplitude of this cosine is set to 0.5 in
order to ensure that the final output is within (-1.0, 1.0).
The final output signal is sent to the bladeRF via an osmocom Sink block, which presents
a hardware-agnostic sink interface.
The sink’s frequency, sample rate, and bandwidth are set to the values described at the
beginning of this section – 464.9375 MHz, 8Msps, and 6 MHz, respectively.
The bladeRF has two Variable Gain Amplifier (VGA) stages in its TX path, as shown in [6].
Generally, TXVGA1 should be increased before TXVGA2. Sliders for these parameters have
been provided in the GUI, via QT GUI Range block. The osmocom Sink block maps its
BB Gain to TXVGA1, and the RF Gain to TXVGA2.
The device-specific arguments are provided via the Device Parameters property string, which
includes the following:
bladerf=0,buffers=128,buflen=8192,transfers=32
The first argument specifies that the first instance of a bladeRF device found on the system
should be used. The remaining items configure properties of the data stream, including
number of sample buffers to use, the size of each sample buffer, and the number of USB
transfers to pre-allocate. Additional buffering ensures that samples are not dropped during
periods of heavy CPU load, with trade-offs of added latency and memory usage.
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An SDR-based FRS Transceiver Nuand, LLC
Receive Path
The design of the RX path is very similar to the TX path, but in reverse. The same sample
rate and bandwidth settings from the TX path are applied. Also, the filter designs from the
TX path are used as a starting point for the associated filter in the RX path, implying that
the decimation rates follow the same interpolation rates.
bladeRF Source
8 Msps
Freq. Xlating FIR
↓4
Decimating FIR
↓5
Polyphase
Filterbank
↓16
•
•
•
Power Squelch NBFM Demod CTCSS Squelch IIR HPF
Audio Sink
25kHz
Power Squelch NBFM Demod CTCSS Squelch IIR HPF
Audio Sink
25kHz
Power Squelch NBFM Demod CTCSS Squelch IIR HPF
Audio Sink
25kHz
Power Squelch NBFM Demod CTCSS Squelch IIR HPF
Audio Sink
25kHz
Figure 9: FRS RX block diagram
Hardware Source and Valve
An osmocom Source block is used to receive samples from the bladeRF. Per previous
calculations, it is tuned to 465.3375 MHz, with a sample rate of 8 Msps, and a bandwidth
of 6 MHz.
In the RX path, the LMS6002D provides a Low Noise Amplifier (LNA) gain stage, and two
VGA gain stages. The LNA stage should be adjusted first, followed by RXVGA1 and then
RXVGA2. The osmocom Source block maps the RF Gain field to the LNA gain, and
the BB Gain to the sum of RXVGA1 and RXVGA2, where RXVGA2 is incremented after
RXVGA1 reaches its max value.
The output of this hardware source is connected to a Valve block allowing all RX processing
to be stopped.
Power Meter
The instantaneous power of the received signal passing through the Valve is filtered via
a Single Pole IIR Filter, converted to dBFs, and then displayed via a QT GUI
Number Sink. This GUI Widget provides a thermometer-style bar that allows the user to
quickly gain a sense of the input power over the full 8 MHz captured at the ADC.
If strong in-band signals cause this GUI item to display a value greater than 0 dBFS, the
input will saturate and likely result in artifacts and distortion. Upon observing the power
meter nearing or exceeding 0dBFS, a user should reduce RX gain values, or introduce external
attenuation if gain values are already at a minimum.
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An SDR-based FRS Transceiver Nuand, LLC
Frequency Translation and Decimating FIR Filters
The signals passed by the Valve are also routed to sections of the flowgraph that perform
frequency translation, filtering, and decimation. Again, like in the TX case, the decima-
tion and filtering is performed in two stages instead of a single stage in order to reduce
computational load.
The first block is a Frequency Xlating FIR Filter, which efficiently translates a
signal to a desired frequency offset, while filtering and decimating.
The Frequency Xlating FIR Filter for FRS channels 1-7 centers the input originally
at -2.7MHz, and the block for channels 8-14 centers the input originally at 2.3 MHz, per
Figure 1.
These blocks are configured to decimate by a factor of 4, yielding an output sample rate of
2 MHz. To avoid aliasing caused by decimation, the FIR filter applied prior to decimation
must reject frequencies outside [-1 MHz, 1 MHz]. Frequencies within [-87.5 kHz, 87.5 kHz]
must be passed, as this region of the spectrum contains the FRS channels. The parameters
used to create a filter that meets these constraints are shown in Table 7.
Table 7: Filter parameters for RX frequency translating and decimating FIR filter
Design Method FIR: Equiripple
Density Factor 20
Order 39
Sampling Frequency (MHz) 8
Passband Frequency (kHz) 21.875
Stopband Frequency (MHz) 1
Passband Weight 1
Stopband Weight 1
Note that although the pass band frequency specified for use with fdatool is 21.875 kHz,
the resulting response of this stage (Figure 10) does not impose any significant attenuation
up through the 87.5 kHz region of interest. In this case, it is found that relaxing the passband
constraint allows fdatool to design a filter with better stop band attenuation.
Next, the 2 MHz signal is passed through a Decimating FIR Filter, which performs
filtering and an additional decimation by a factor of 5, resulting in a 400 kHz signal. This
implies that frequencies outside of [-200 kHz, 200 kHz] must be rejected, while ensuring the
FRS content in [-87.5 kHz, 87.5 kHz] is passed without attenuation. Table 8 presents the
filter design parameters.
As shown in Figure 10, this filter does indeed meet the desired requirements, as does the
entire response of the two filtering stages.
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An SDR-based FRS Transceiver Nuand, LLC
Table 8: Filter parameters for RX decimating FIR filter
Design Method FIR: Equiripple
Density Factor 20
Order 39
Sampling Frequency (MHz) 2
Passband Frequency (kHz) 87.5
Stopband Frequency (kHz) 200
Passband Weight 1
Stopband Weight 1
Frequency (MHz)
0 0.5 1 1.5 2 2.5 3 3.5
Magnitude (dB)
-200
-150
-100
-50
0
Combined Response
First (4x) Stage
Second (5x) Stage
Figure 10: Magnitude response of RX decimating FIR filters
Polyphase Channelizer
In this flowgraph, the usage of the Polyphase Channelizer is very similar to that of
the Polyphase Synthesizer in the TX path.
The input to each channelizer is a 400 kHz wide signal, centered on FRS channels 4 or 11
(see Figure 6). By configuring the channelizer for 16 channels and using the same channel
mapping presented in Section 4.1.4, indices 0-6 are associated with FRS channels 1-7 or 8-14.
Each of the outputs containing a single FRS channel are routed to a Valve block. Similar
to the Mute blocks used in the TX path, these are controlled by GUI checkboxes to allow
the user to select which FRS channels to listen to. However, the Valve is used to disable
downstream blocks when opened. This eliminates the unnecessary processing of channels
that the user is not interested in.
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An SDR-based FRS Transceiver Nuand, LLC
Bus ports [9] are used to route the remaining 9 unused outputs to a single Null Sink,
rather than having 9 separate Null Sinks displayed on the flowgraph; this is merely a
matter of preference to reduce visual clutter in the flowgraph.
Squelch and FM Demodulation
A Power Squelch block is placed before the NBFM Receive block to prevent the CPU
from performing all the demodulations simultaneously when no active signal is being received.
The Gate option is set to “Yes,” meaning that when the incoming signal does not exceed
a specified power threshold, no samples will be passed to the output causing downstream
blocks to stop. This is intended to further eliminate unnecessary processing.
The squelch level is controllable by the user via a slider provided by a QT GUI Range
block. The required level for this block is expected to vary with the operating environment.
A fairly high squelch threshold default has been selected for to avoid immediately passing
noise when the flowgraph is started.
A value of 0.0125 for the alpha parameter of the Power Squelch is selected to signif-
icantly favor past samples in order to avoid breaking squelch due to any transient spikes.
This parameter can be tuned in the flowgraph via an rx power squelch alpha.
After breaking squelch, a signal then enters the NBFM Receive block. No decimation is
necessary
5
, so the Audio Rate (output sample rate) parameter is set equal to the Quadrature
Rate (input sample rate).
The Max Deviation parameter of 2.5 kHz is derived from the FRS specification [2].
The NBFM Receive block is designed to perform de-emphasis using an Infinite Impulse
Response (IIR) filter, however, de-emphasis is not needed in this design. Therefore a value
for the RC time constant tau parameter is selected such that the resulting filter response is
flat up to at least the 3.125 kHz max audio frequency.
The IIR filter taps (in transfer function from) are calculated by the NBFM Receive block
as follows
6
:
5
Decimation by a factor of two here could be performed to further reduce computational load.
6
As found in fm emph.py as of GNU Radio commit ee369b92
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An SDR-based FRS Transceiver Nuand, LLC
ω = tan
1
τ
F
s
× 2
a
0
= 1 b
0
=
ω
1 + ω
a
1
=
ω − 1
ω + 1
b
1
= b
0
taps
feed-back
= [a
0
, a
1
] taps
feed-forward
= [b
0
, b
1
]
Given the F
s
= 25kHZ sample rate, MATLAB’s fvtool can be used to view the response
for various values of τ . A value of τ = 5 × 10
−6
is found to yield a stable filter with a
sufficiently flat response through the frequencies up to 3.125 kHz FRS maximum, as shown
in Figure 11
Frequency (kHz)
0 2 4 6 8 10 12
Magnitude (dB)
-60
-50
-40
-30
-20
-10
0
Magnitude Response (dB)
Figure 11: Magnitude response of NBFM de-emphasis filter
As of GNU Radio commit ee369b92, the NBFM Receive block also performs a LPF
operation to limit the audio output. The limit is hard coded at 2.7 kHz with a 0.5 kHz
transition band using a Hamming window.
Following the FM demodulation, the audio signal is routed to both a CTCSS Squelch
block and directly to a Selector block. When the GUI dropdown for the desired CTCSS
tone is set to “None” the Selector passes the output of the NBFM Receive block to the
next stage. Otherwise, the Selector passes the output of the CTCSS Squelch block,
which is configured to pass samples when the CTCSS tone associated with the current GUI
dropdown widget selection is detected.
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An SDR-based FRS Transceiver Nuand, LLC
Adequate values for the CTCSS Squelch values are determined experimentally by inter-
facing with handheld FRS radios, such as those described in Section 5. A Length value of
6250 samples (250 ms at 25 kHz) and Level of 0.010 are found to provide desirable squelch
functionality.
CTCSS Tone IIR Filter
An astute listener can hear a number of the higher-frequency CTCSS tones. Therefore,
a high-pass filter is needed to reject audio below 250.3 Hz. At a 25 kHz sample rate, an
acceptable FIR filter is found to require a large number of taps
7
. Instead, an IIR filter is
designed using fdatool, as this is expected to yield significantly fewer taps. The design
parameters are shown in Table 9, and the filter response is presented in Figure 12.
Table 9: Filter parameters of RX CTCSS-blocking high pass filter
Design Method IIR: Elliptic
Option: Match Exactly both (pass and stop)
Order 7
Sampling Frequency (kHz) 25
Stopband Frequency (Hz) 250.3
Passband Frequency (Hz) 300
Passband Attenuation (dB) 1
Stopband Attenuation (dB) 50
Audio Sinks
The audio from each FRS channel is provided to the system via the Audio Sink block.
As done in the TX path, “pulse” is specified for the Device Name parameter to utilize the
PulseAudio subsystem. It is possible to use one sink per channel because the underlying
audio subsystem handles multiple inputs (from each sink). When multiple channels are
being received, all of them will be heard and combined by the PulseAudio subsystem.
7
Approximately 900 taps for a FIR: Equiripple and 50dB of stopband attenuation.
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An SDR-based FRS Transceiver Nuand, LLC
Frequency (kHz)
0 2 4 6 8 10 12
Magnitude (dB)
-70
-60
-50
-40
-30
-20
-10
0
Frequency (kHz)
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8
Magnitude (dB)
-70
-60
-50
-40
-30
-20
-10
0
Frequency: 0.3005981
Magnitude: -0.5999716
Frequency: 0.2723694
Magnitude: -50.27843
Figure 12: Response of RX CTCSS-blocking high pass filter
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An SDR-based FRS Transceiver Nuand, LLC
Testing and Evaluation
Flowgraph GUI
As noted in previous sections, various blocks provide GUI widgets that may be used to
control and interact with the running flowgraph. Screen captures of this GUI are shown in
Figures 13 and 14.
The top of the GUI provides quick access to CTCSS tone selection and the ability to en-
able/disable the RX and TX functionality.
By default, the CTCSS dropdown is set to “None.” This can be changed at any time, but it
is recommended to make changes while RX and TX are disabled, as portions of the flowgraph
will be reconfigured.
The “Enable RX” checkbox is a master enable for all reception. Two master enable options
are provided for TX. A “Continuous Enable” checkbox enables the TX front end while the
checkbox is marked. The “Push to Talk” button enables the TX front end only while this
button is pressed.
The remainder of the GUI is split between an RX tab and a TX tab. Both views provide
checkboxes for each FRS channel. Marking the checkbox enables reception/transmission of
the associated channel. Gain controls are also provided in each view.
The RX view also provides control over the Power Squelch block as the noise floor may
vary in different environments.
Waterfall plots are provided for both groups of channels (1-7 and 8-14). These plots display
received signals in the frequency domain prior to any squelch blocks. An “Input Power”
meter is placed between these plots to provide a quick means to determine if gains are set
too high or too low.
The TX side also provides waterfall plots. These are intended to provide feedback in response
to the user’s checkbox selections.
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An SDR-based FRS Transceiver Nuand, LLC
Figure 13: RX Section of the flowgraph GUI
Figure 14: TX Section of the flowgraph GUI
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An SDR-based FRS Transceiver Nuand, LLC
Nuand bladeRF SDR
The Nuand bladeRF is a USB 3.0 SDR with a frequency range of 300 MHz to 3.8 GHz,
full-duplex capabilities, a maximum bandwidth of 28 MHz, 12-bit samples, and a maximum
sample rate of 40 MHz [10]. The architecture of the platform is illustrated in Figure 15.
The bladeRF utilizes a Cypress FX3 USB3 device controller, which allows data to be trans-
ferred between its USB 3.0 SuperSpeed connection and its parallel GPIF II interface (run-
ning half-duplex at a speed of 100MHz). This is connected to an Altera Cyclone IV E Field
Programmable Gate Array (FPGA). The FPGA provides buffering and additional signal
processing, as well as control of the Lime Microsystems LMS6002D Field-Programmable RF
Transceiver.
The LMS6002D transceiver IC, whose architecture is depicted in in Figure 16, has multiple
input and output RF ports which are tuned to different frequency ranges. The architecture
is split into a low band (300 MHz - 1500 MHz) and high band (1500 MHz - 3800 MHz). The
LMS6002D contains independently configurable RX and TX signal chains, which allows the
bladeRF platform to provide full-duplex capabilities. Also note that the RX and TX mixers
each have their own PLLs, which synthesize desired tuning frequencies from a common
input clock. These are what allow the device to operate the RX and TX paths at different
frequencies.
A single VCTCXO running at 38.4MHz supplies the FX3 and Silicon Labs Si5338 low-jitter
clock generator. All sample rate clocks are generated by the Si5338 with high accuracy,
allowing for flexible and arbitrary clocking of the ADC/DAC. The Si5338 also supplies clocks
to the Altera Cyclone FPGA and to the Lime Microsystems LMS6002D.
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An SDR-based FRS Transceiver Nuand, LLC
Figure 15: Nuand bladeRF architecture
Figure 16: Lime Microsystems LMS6002D architecture
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An SDR-based FRS Transceiver Nuand, LLC
Reception
To initially test reception, an Agilent E4433B signal generator [11] is used to supply FM
modulated tones to the bladeRF. This device is configured to supply an FM-modulated 600
Hz tone with a deviation of 2.5 kHz and an amplitude of -60 dBm.
The channel mapping is verified by supplying a signal at a specific FRS channel and checking
that the demodulated signal is played only when that associated channel is enabled in the
GUI. This is repeated for each of the 14 FRS channels.
The per-channel filters are evaluated in a similar fashion, by verifying that supplying a
signal to the bladeRF on one channel does not yield significant interference on neighboring
channels.
Next, it is verified that none of the channels are attenuated significantly more than others, as
one might expect to occur if the decimating FIR filters had been designed to be too narrow.
This is done by sweeping across 200 kHz of bandwidth with the signal generator, and noting
the signal level in the waterfall plots provided in the designed FRS Transceiver GUI. As
expected, attenuation begins occurring before channels 1 and 8, and after channels 7 and
14. No significant attenuation is observed with the regions containing the groups of 7 FRS
channels.
Figure 17: Cobra CXT545 FRS handheld transceiver used to test flowgraph
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An SDR-based FRS Transceiver Nuand, LLC
Next, interoperability with commercially-available handheld FRS transceivers is tested. A
pair of Cobra CXT545 [12] walkie talkies, configured for low power operation on FRS channels
(Figure 17) are used. These devices support all 38 of the commonly used CTCSS tones listed
in Table 2, with 67.0 Hz being CTCSS tone #1, and 250.3 Hz being tone #38.
Within the confines of a lab environment, the output power of these radios is found to
saturate the bladeRF input. This is addressed by reducing the gain values used for reception
and/or using external attenuators.
The channel mapping and per-channel filter tests are performed again using these devices.
The CTCSS squelch functionality is verified by transmitting with each tone on the handheld,
and confirming that the audio is played by the flowgraph only when “None” or the associated
CTCSS tone is selected in the GUI.
Transmission
The waterfall plots in the TX portion of the flowgraph are used to provide confirmation that
each channel’s signal is located at the correct offset. To further evaluate the signal being
supplied to the bladeRF for transmission, a Qt GUI Frequency Sink is connected in
parallel to the osmocom Sink block. For a higher-resolution view of a group of channels,
a frequency sink block is placed in parallel with the first Interpolating FIR Filter
block.
Figure 18 depicts the digital representation of the signal containing a concurrent transmission
on channels 8 through 14 while Figure 19 show a transmission containing only channel 8. A
plot containing the simultaneous transmission on all 14 channels is presented in Figure 20.
Figure 18: FFT plot of digital signal containing FRS channels 8-14
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An SDR-based FRS Transceiver Nuand, LLC
Figure 19: FFT plot of digital signal containing FRS channel 8
Figure 20: FFT plot of digital signal containing all FRS channels
Per these plots, the channels are located at the expected frequencies, with bandwidths within
the 12.kHz maximum. The expected aliases that occur as a side effect of polyphase synthesis
[8] are significantly rejected by the designed filters and are observed to occur 100 dB below
the signals of interest.
Next, the output of the bladeRF’s TX port is observed, using an Agilent E4406A Vector
Signal Analyzer (VSA) [13]. Again, the locations and widths channels are first verified to be
correct.
The signal levels of the FRS signals and the expected aliases are found to be within acceptable
limits. It is noted that the magnitudes do differ from the digital representation to some
degree. This may be attributed to quantization error that is inherent in converting a digital
floating point signal to an analog signal through the hardware’s 12-bit Digital to Analog
Converter (DAC). This can be simulated in GRC by adding a Quantizer block (found
under “Impairment Models”) prior to the osmocom Sink block.
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An SDR-based FRS Transceiver Nuand, LLC
As discussed in Section 4, the bladeRF’s TX frequency is tuned 200 kHz under 465.1375
MHz – the frequency that is equidistant from both groups of channels – to avoid distortion
caused by any IQ imbalance.
The effectiveness of this strategy is made clear by Figure 22. Note that to the left of each
group of FRS signals, a significantly attenuated image from the other group is present. By
ensuring these images do not overlap with the intended signal, a higher Signal-to-Noise and
Distoration ratio (SINAD) is achieved. For a laboratory experiment, 40 dB of rejection for
these images and the carrier caused by residual DC offset is certainly acceptable. Possible
approaches to further reduce the undesired spectral content is to utilize more aggressive
digital filters (likely requiring a large number of taps), external filters, or a combination of
both.
Figure 21: FFT plot of analog signal containing FRS channels 1-7
Figure 22: FFT plot of analog signal containing all 14 FRS channels
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An SDR-based FRS Transceiver Nuand, LLC
Software and System Configuration
The design and evaluation of the flowgraph described in this document is performed with
XUbuntu 14.10 [14], on machines with Intel i7 processors, more than 4 GB of RAM, and
either Intel XHCI 7 Series or Renesas uPD720202 USB 3.0 host controllers. Similar or better
systems are recommended.
Due to the sample rates used in this flowgraph, USB 3.0 is required for full-duplex operation.
Be aware that even for half-duplex operation, USB 2.0 controllers may not be able to support
the required 32 MB/s
8
Table 10 lists the minimum required software versions needed to use this transceiver. Newer
versions of the listed items should also be compatible.
Table 10: Minimum required component versions
Component Version
bladeRF FPGA 0.1.2
bladeRF Firmware 1.8.0
libbladeRF 1.2.1
GNU Radio 3.7.7
gr-osmosdr 0.1.4-g48045b59
The “Getting Started” guide on the bladeRF wiki [15] provides instructions for installing
the latest versions of the above items. Those new to the bladeRF and GNU Radio may wish
to use the GNU Radio Live SDR Environment [16] , an Ubuntu-based live image with GNU
Radio and SDR support pre-installed.
Conclusions
This paper has demonstrated a successful design and implementation of a prototype FRS ra-
dio system in a manner that takes advantage of the features provided by the Nuand bladeRF
SDR platform. In doing so, it has exemplified topics including performance-conscious filter
design and operation on multiple narrow-band channels via GNU Radio’s Polyphase Chan-
nelizer/Synthesizer blocks. Furthermore, constraints and challenges presented by real-world
nonidealities have been addressed throughout the design, implementation, and evaluation
processes. Upon developing a level of comfort with these topics, readers should be able to
experiment with, modify, and further extend the GNU Radio flowgraph presented in this
work, as well as apply these concepts to new designs.
8
Although the theoretical max throughput of USB 2.0 is 60 MB/s, the practical achievable throughput
is significantly less due to protocol overheads.
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An SDR-based FRS Transceiver Nuand, LLC
Flowgraph Diagrams
Figure 23: Full FRS transceiver flowgraph
30
An SDR-based FRS Transceiver Nuand, LLC
Figure 24: RX path of FRS transceiver
31
An SDR-based FRS Transceiver Nuand, LLC
Figure 25: TX path of FRS transceiver, slightly re-arranged to fit page
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An SDR-based FRS Transceiver Nuand, LLC
References
[1] Family Radio Service (FRS).
https://www.fcc.gov/encyclopedia/family-radio-service-frs. Accessed: 2015-03-02.
[2] eCRF – Code of Federal Regulations, Part 95 - Personal Radio Services, Subpart B - Family Radio
Service (FRS). http://www.ecfr.gov/cgi-bin/text-idx?SID=
5ddd3bab07204521939c4b1580b7a692&node=pt47.5.95&rgn=div5#sp47.5.95.b.
Accessed: 2015-03-02.
[3] GNU Radio. http://gnuradio.org/. Accessed: 2015-03-02.
[4] Nuand bladeRF. https://www.nuand.com/bladeRF. Accessed: 2015-03-02.
[5] Field Programmable RF ICs: LMS6002D.
http://www.limemicro.com/products/field-programmable-rf-ics-lms6002d.
Accessed: 2015-03-09.
[6] LMS6002 Datasheet (1.1.0).
http://www.limemicro.com/download/LMS6002Dr2-DataSheet-1.2r0.pdf. Accessed:
2015-03-09.
[7] PulseAudio. http://www.freedesktop.org/wiki/Software/PulseAudio. Accessed:
2015-03-03.
[8] PFB Channelizers and Synthesizers. http://www.trondeau.com/examples/2014/1/23/
pfb-channelizers-and-synthesizers.html. Accessed: 2015-03-02.
[9] Working with GRC Busports.
http://www.trondeau.com/blog/2014/2/27/working-with-grc-busports.html.
Accessed: 2015-03-10.
[10] Nuand bladeRF Product Brief. https://www.nuand.com/bladeRF-brief.pdf. Accessed:
2015-10-24.
[11] Keysight E4433B ESG-D Series Digital RF Signal Generator.
http://www.keysight.com/en/pd-1000002824%3Aepsg%3Apro-pn-E4433B/
esg-d-series-digital-rf-signal-generator-4-ghz?cc=US&lc=eng. Accessed:
2015-03-10.
[12] Cobra CXT545. https://www.cobra.com/products/max-performance/cxt-545. Accessed:
2015-03-10.
[13] Keysight E4406A VSA Transmitter Tester. http://www.keysight.com/en/pd-1000002790%
3Aepsg%3Apro-pn-E4406A/vsa-transmitter-tester-7-mhz-to-4-ghz?cc=US&lc=eng.
Accessed: 2015-03-10.
[14] XUbuntu 14.10 released! http://xubuntu.org/news/14-10-release/. Accessed: 2015-03-11.
[15] bladeRF ”Getting Started” guide for Linux.
https://github.com/Nuand/bladeRF/wiki/Getting-Started:-Linux. Accessed:
2015-10-24.
[16] GNU Radio Live SDR Environment.
https://gnuradio.org/redmine/projects/gnuradio/wiki/GNURadioLiveDVD.
Accessed: 2015-10-24.
33